Method and apparatus for detecting spreading-code synchronization for spectrum spreading signals

ABSTRACT

A spreading-code synchronization detection method for a spectrum spreading signal that increases the sensitivity of spreading-code synchronization detection and in which a spreading-code synchronization detection is applied to a spectrum spreading signal obtained by spectrum-spreading data, having a bit transition period equal to a multiple of one period of a spreading code, with the spreading code. A process for obtaining a linear-addition correlation-calculation result equal to a value obtained by linear additions of the results of correlation calculations between the spectrum spreading signal and the spreading code is performed every unit period, which is a multiple of one period of the spreading code and shorter than the bit transition-period. The absolute value of the linear-addition correlation-calculation result obtained every unit period is calculated. The absolute value of the linear-addition correlation-calculation result obtained every unit period is added for a plurality of unit periods. A correlation point is detected from a value obtained by adding the absolute values.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.10/380,650, filed on Mar. 14, 2003, the disclosure of which isincorporated herein by reference.

TECHNICAL FIELD

The present invention relates to methods and apparatuses for detectingspreading-code synchronization for spectrum spreading signals, such asGPS (global positioning system) satellite signals.

BACKGROUND ART

In a GPS system for measuring the position of a mobile body by usingartificial satellites (GPS satellites), a GPS receiver has a basicfunction of receiving signals from four or more GPS satellites,calculating the position of the receiver from the received signals, andinforming the user of the position.

The GPS receiver demodulates the signals sent from the GPS satellites toacquire the orbit data of the GPS satellites, and uses simultaneousequations to calculate the three-dimensional position of the receiverfrom the orbits and time information of the GPS satellites, and thedelay time of the received signals. The reason why four GPS satellitesfrom which signals are received are required is to remove the effect ofan error between the time used in the GPS receiver and the time used bythe satellites.

A commercially available GPS receiver receives a spectrum spreadingsignal radio wave called a C/A (clear and aquisition) code in an L1 bandfrom a GPS satellite (Navstar) to perform calculations for positionmeasurement.

The C/A code is a PN (pseudorandom noise) series code having atransmission-signal speed (chip rate) of 1.023 MHz and a code length of1023, such as a Gold code, and is a signal obtained by BPSK (binaryphase shift keying) modulating a carrier wave (hereinafter called acarrier) having a frequency of 1575.42 MHz by a signal obtained byspreading data of 50 bps. In this case, since the code length is 1023, aPN-series code is repeated in the C/A code with 1023 chips being used asone period (therefore, one period is equal to 1 millisecond), as shownin FIG. 20(A).

The PN-series code in the C/A code differs in each GPS satellite. TheGPS receiver can detect in advance a PN-series code used by each GPSsatellite. In addition, the GPS receiver understands from a navigationmessage like that described later whether the receiver can receive asignal from each GPS satellite at its position at the point of time.Therefore, for three-dimensional position measurement, for example, theGPS receiver receives radio waves which can be obtained at its positionat the point of time from four or more GPS satellites, applies inversespectrum spreading to the radio waves, and performs calculations forposition measurement to obtain its position.

As shown in FIG. 20(B), one bit of satellite-signal data is transferredin units of 20 periods of the PN-series code, that is, in units of 20milliseconds. In other words, the data transmission rate is 50 bps. ThePN-series code in one period, that is, 1023 chips, is inverted betweenwhen the corresponding bit is “1” and when the bit is “0.”

As shown in FIG. 20(C), one word is formed of 30 bits (600 milliseconds)in the GPS, and one sub-frame (six seconds) is formed of 10 words asshown in FIG. 20(D). In the first word of a sub-frame always includes apreamble having a fixed bit pattern even when data is updated, and datais transferred after the preamble.

Further, one frame (30 seconds) is formed of five subframes. Thenavigation message is transferred in units of one-frame data. Firstthree sub-frames in one-frame data includes information unique to asatellite, called ephemeris information. The information includes aparameter used for obtaining the orbit of the satellite, and the timewhen the satellite sent the signal.

All GPS satellites have an atomic clock, and use common timeinformation. The time when a GPS satellite sends a signal is indicatedin units of seconds of the atomic clock. The PN-series code of a GPSsatellite is generated in synchronization with the atomic clock.

Orbit information in the ephemeris information is updated in units ofseveral hours. Until an update is performed, the same information isused. The orbit information in the ephemeris information can be storedin a memory of the GPS receiver so as to use the same informationprecisely for the several hours. The time when a GPS satellite sends asignal is updated in units of seconds.

The navigation message included in the remaining two sub-frames in theone-frame data is information sent in common from all satellites, calledalmanac information. The almanac information is transferred in 25frames, and includes rough-position information of each GPS satelliteand information indicating which GPS satellite is available. The almanacinformation is updated in units of several months. Until an update isperformed, the same information is used. The almanac information can bestored in a memory of the GPS receiver so as to use the same informationprecisely for the several months.

To receive a signal from a desired GPS satellite to obtain theabove-described data, the carrier is first removed, the signal sent fromthe GPS satellite is phase-synchronized with the C/A code by using thesame PN-series spreading code as the C/A code used by the GPS satellite,prepared in the GPS receiver to capture the signal, and inverse spectrumspreading is performed. When phase synchronization with the C/A code isobtained and inverse spreading is performed, each bit is detected, and anavigation message, including time information, can be obtained from thesignal sent from the GPS satellite.

The signal sent from the GPS satellite is captured by C/A-code phasesynchronization search. In the phase synchronization search, correlationbetween the spreading code of the GPS receiver and the spreading code ofthe signal received from the GPS satellite is detected, and when acorrelation value obtained as the result of correlation detection islarger than a value specified in advance, for example, it is determinedthat both codes are synchronized. When it is determined that they arenot synchronized, the phase of the spreading code of the GPS receiver iscontrolled by some synchronization method to synchronize the spreadingcode of the GPS receiver with the spreading code of the received signal.

Since a GPS satellite signal is obtained by BPSK modulating the carrierby a signal obtained by spreading data with a spreading code, asdescribed above, the carrier and the data need to be synchronized inaddition to the spreading code when the GPS receiver receives the GPSsatellite signal. The synchronization of the spreading code and that ofthe carrier cannot be performed independently.

The GPS receiver usually converts the carrier frequency of the receivedsignal to an intermediate frequency several megahertz from the carrierfrequency, and performs the above-described synchronization detectionprocess with a signal having the intermediate frequency. The carrier inthe intermediate-frequency signal mainly includes a frequency errorcaused by a Doppler shift corresponding to the moving speed of the GPSsatellite and an error in the frequency of a local oscillator, generatedinside the GPS receiver.

Therefore, the carrier frequency in the intermediate-frequency signal isunknown due to these frequency-error factors, and the carrier frequencyneeds to be searched for. A synchronization point (synchronized phase)in one period of the spreading code depends on the positionalrelationship between the GPS receiver and the GPS satellite. Since thispositional relationship is also unknown, some synchronization method isrequired, as described above.

Conventional GPS receivers use frequency search for the carrier and aspreading-code synchronization detection method which uses a slidingcorrelator, a DLL (delay locked loop, and a costas loop. A descriptionthereof will be given below.

Usually a reference-frequency oscillator provided for the GPS receiversis scaled down to generate a clock signal used for driving a PN-codegenerator in the GPS receivers. As the reference-frequency oscillator, ahigh-precision crystal oscillator is used, and a local oscillatingsignal used for converting a signal received from a GPS satellite to anintermediate-frequency signal is generated from the output of thereference-frequency oscillator.

FIG. 21 is a view used for describing the frequency search.Specifically, when the frequency of the clock signal used for drivingthe spreading-code generator in the GPS receiver is a frequency f1,phase synchronization search is performed for the spreading code, inother words, the phase of the spreading code is sequentially shifted byone chip, correlation between a GPS received signal and the spreadingcode is detected at each chip phase, and a correlation peak is detectedto detect a phase at which synchronization is acquired.

When the clock signal has the frequency f1, if there is no phase atwhich synchronization is acquired, among 1023 chip phases throughsearch, a scaling ratio for the reference-frequency oscillator ischanged, for example, to change the frequency of the driving clocksignal to a frequency f2, and phase search is performed in the same wayfor 1023 chips. As shown in FIG. 21, the frequency of the driving clocksignal is changed step by step and phase search is repeated. Theabove-described operation is called frequency search.

When the frequency of the driving clock signal, for whichsynchronization can be acquired is detected through the frequencysearch, a final spreading-code phase synchronization detection isperformed at the clock frequency. With this, even if the oscillationfrequency of the crystal frequency oscillator is shifted, a satellitesignal can be captured.

When the above-described conventional method is used as a spreading-codesynchronization detection method, however, it is, in principle, notsuited to high-speed synchronization, and to compensate this, a receiverneeds to have multiple channels and to search for a synchronizationpoint in parallel. When the synchronization of the spreading code andthat of the carrier require time as described above, the GPS receiverhas a slow response and causes inconvenience in use.

Due to the improvement of the capability of hardware, typical of whichis DSPs (digital signal processors), without using a sliding correlationmethod like that described above, a method for performing spreading-codephase synchronization detection at a high speed is implemented by usinga digital matched filter.

There have been known digital matched filters using a transversal filteror fast Fourier transform (hereinafter called FFT). Usually, a digitalmatched filter performs processing in units of periods of the spreadingcode.

When spreading-code synchronization detection is performed only with theresults of correlation calculations for one period of the spreadingcode, a detection sensitivity is low. Therefore, to increase thedetection sensitivity, a method has been conventionally used, in whichthe sum of the square of the result of correlation calculations for oneperiod of the spreading code is accumulated. According to this method,since a correlation value at a correlation point is made larger thancorrelation Values at non-correlation points irrespective of thepositive and negative polarities of correlation values, the detectionsensitivity is increased.

Since a noise component is also accumulated without being offset in themethod in which the sum of the squares is accumulated, however, a losscaused by square operations is large and the degree of improvement ofthe detection sensitivity is low in a receiving condition having a lowC/N (carrier-to-noise ratio).

There can be another method in which, not the sum of the squares, butthe linear sum of the result of correlation calculations for one periodof the spreading code is accumulated. In the linear sum, noisedistributed at random is offset and reduced.

In a GPS signal, a spectrum spreading signal includes 50-bps navigationdata, and a bit transition period is set to 20 times (20 milliseconds)the period (one millisecond) of the spreading code, as shown in FIG. 20.Therefore, when the linear sum of the result of correlation calculationsfor one period of the spreading code is accumulated for 20 or moremilliseconds, since a correlation value has a inverted polarity fromwhen a bit transition occurs, and offset, the accumulated value becomessmall. The linear sum cannot be accumulated simply.

The present invention has been made in consideration of the foregoingpoints. An object of the present invention is to allow the sensitivityof spreading-code synchronization detection for a spectrum spreadingsignal of data having a bit transition period which is a multiple of oneperiod of the spreading code, such as the above-described GPS signal, tobe greatly improved.

DISCLOSURE OF INVENTION

To solve the foregoing issues, a spreading-code synchronizationdetection method for a spectrum spreading signal according to thepresent invention (1) is

a method for spreading-code synchronization detection for a spectrumspreading signal obtained by spectrum-spreading data having a bittransition period which is a multiple of one period of a spreading codewith the spreading code, characterized by comprising:

unit-period correlation-calculation linear-addition means for performinga process for obtaining a liner-addition correlation-calculation resultequal to a value obtained by linear additions of the results ofcorrelation calculations between the spectrum spreading signal and thespreading code, every unit period which is a multiple of one period ofthe spreading code and shorter than the bit transition period;

an absolute-value calculation step of calculating the absolute value ofthe liner-addition correlation-calculation result obtained every unitperiod in the unit-period correlation-calculation linear-addition step;

an absolute-value addition step of adding the absolute value of theliner-addition correlation-calculation result obtained every unitperiod, obtained in the absolute-value calculation step, for a pluralityof unit periods; and

a correlation-point detection step of detecting a correlation point froma value obtained by adding the absolute values in the absolute-valueaddition step.

In the present invention (1) having the above-described structure, notthe square sum but the linear sum is obtained in a unit period shorterthan the bit transition period of the data. The sum of the absolutevalue of the linear sum in the unit period is accumulated for aplurality of unit periods, and a correlation point is detected from theaccumulated sum of the absolute values.

In this case, since where a bit transition occurs in the data isunknown, positive and negative correlation values are offset by eachother in a process for calculating the linear sum in the unit period,and especially when a bit transition occurs at the center of a unitperiod, since the positive and negative correlation values are offset,the linear sum of the calculated correlation values becomes zero.

In the present invention (1), however, since the unit period is setshorter than the bit transition period, when a bit transition occurs atthe center of a unit period, a bit transition never occurs at the centerof a unit period immediately before or after the unit period.

Like the present invention (2), for example, when the unit period ishalf the bit transition period, if a bit transition occurs at the centerof a unit period, the unit periods immediately before and after the unitperiod do not include a bit transition. Therefore, in the unit periodsimmediately before and after, the linear sum of the results ofcorrelation calculations, which do not receive any effect of offsetscaused by bit transitions is obtained. When a correlation point isdetected by the sum of the absolute values thereof, it is expected thatthe detection sensitivity is increased.

A spreading-code synchronization detection method for a spectrumspreading signal according to the present invention (3) is

a spreading-code synchronization detection method for a spectrumspreading signal obtained by spectrum-spreading data having a bittransition period which is a multiple of one period of a spreading codewith the spreading code, characterized by comprising:

a unit-period correlation-calculation linear-addition step of obtaininga first liner-addition correlation-calculation result equal to a valueobtained by linear additions of the results of correlation calculationsbetween the spectrum spreading signal and the spreading code, every unitperiod which is a multiple of one period of the spreading code and equalto or shorter than the bit transition period, and

of dividing the unit period into two periods, a first-half period and asecond-half period, and of obtaining every unit period a secondliner-addition correlation-calculation result equal to the sum of afirst linear sum equal to the linear sum of the results of correlationcalculations between the spectrum spreading signal and the spreadingcode in one of the first-half period and the second-half period and asecond linear sum equal to the linear sum of the results of correlationcalculations between the spectrum spreading signal and the spreadingcode in a state in which one of the spectrum spreading signal and thespreading code is inverted in sign, in the other one of the first-halfperiod and the second-half period;

an absolute-value addition step of adding the sum of the absolute valueof the first liner-addition correlation-calculation result and theabsolute value of the second liner-addition correlation-calculationresult for a plurality of unit periods; and

a correlation-point detection step of detecting a correlation point froma value obtained by adding the absolute values in the absolute-valueaddition step.

In the present invention (3) having the above-described structure, thesum of the absolute value of the first liner-additioncorrelation-calculation result and the absolute value of the secondliner-addition correlation-calculation result is always constantirrespective of the phase relationship between the unit period and thebit transition position. Therefore, when the sum of the absolute valuesis accumulated for a plurality of unit periods in the absolute-valueaddition step, the accumulated value is the same number of times the sumof the absolute values as the number of the plurality of unit periods.

Therefore, it becomes easier to set a threshold used for detecting acorrelation point in the correlation-point detection step, and thedetection sensitivity is also improved.

The present invention (4) is characterized in a spreading-codesynchronization detection method for a spectrum spreading signaldescribed in the present invention (3), in that the unit period is setequal to the bit transition period, and

a phase shift between the unit period and the bit transition position isestimated by the ratio between the first linear-additioncorrelation-calculation result and the second linear-additioncorrelation-calculation result, and the phase shift between the unitperiod and the bit transition position is compensated for by theestimated phase shift.

In the present invention (4), the unit period is set equal to the bittransition period, and a phase shift between the unit period and the bittransition position is estimated by the ratio between the firstlinear-addition correlation-calculation result and the secondlinear-addition correlation-calculation result. The phase shift betweenthe unit period of the spectrum spreading signal and the bit transitionposition of the data is compensated for by the estimated phase shift.With this, since a bit transition does not occur during a unit period,the linear addition of correlation values in a unit period alwaysindicates the maximum value, and the detection sensitivity ofspreading-code synchronization is further improved.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a view used for describing a main operation in aspreading-code synchronization detection method for a spectrum spreadingsignal according to a first embodiment of the present invention.

FIG. 2 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to the firstembodiment of the present invention.

FIG. 3 is a view showing an example spectrum of a correlation detectionoutput.

FIG. 4 is a block diagram showing an example structure of a digitalmatched filter used in an embodiment of the present invention.

FIG. 5 is a block diagram showing another example structure of a digitalmatched filter used in an embodiment of the present invention.

FIG. 6 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to a secondembodiment of the present invention.

FIG. 7 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to a thirdembodiment of the present invention.

FIG. 8 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to a fourthembodiment of the present invention.

FIG. 9 is a view used for describing the operation of a main section inthe fourth embodiment.

FIG. 10A and FIG. 10B are views used for describing the main section inthe fourth embodiment.

FIG. 11 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to a fifthembodiment of the present invention.

FIG. 12 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to a sixthembodiment of the present invention.

FIG. 13 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to a seventhembodiment of the present invention.

FIG. 14 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to an eighthembodiment of the present invention.

FIG. 15 is a view used for describing the operation of a main section inthe eighth embodiment.

FIG. 16 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to a ninthembodiment of the present invention.

FIG. 17 is a view used for describing a main section in the ninthembodiment.

FIG. 18 is a flowchart used for describing a processing flow in theninth embodiment.

FIG. 19 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to a tenthembodiment of the present invention.

FIG. 20 is a view showing the structure of a signal sent from a GPSsatellite.

FIG. 21 is a view used for describing a conventional carrier andconventional spreading-code synchronization processing.

FIG. 22 is a view used for describing an embodiment of the presentinvention.

BEST MODE FOR CARRYING OUT THE INVENTION

A case in which a method for detecting spreading-code synchronizationfor spectrum spreading signals according to an embodiment of the presentinvention is applied to spreading-code synchronization detection of aGPS signal in the above-described GPS receiver will be described belowby referring to the drawings.

First Embodiment

FIG. 2 is a block diagram showing an example structure of aspreading-code synchronization detection section in the GPS receiver,the section serving as a spreading-code synchronization detectionapparatus for spectrum spreading signals according to a firstembodiment. In FIG. 2, a received signal r(n) is anintermediate-frequency signal in which the carrier of a signal(spectrum-spreading signal) sent from a GPS satellite and received by aGPS antenna, not shown, has been low-frequency-converted to anintermediate frequency of 1.023 MHz.

To make the description simple, it is assumed that carriersynchronization is obtained for the received signal r(n) in theembodiment shown in FIG. 2. It is actually necessary to search for thecarrier frequency by a method, such as that described later, and toobtain carrier synchronization.

The received signal r(n) is converted by an A/D converter 1 to a digitalsignal and then sent to a digital matched filter 2. A spreading code ofone period is sent from a spreading-code generation section 3 to thedigital matched filter 2. At this moment, the spreading-code generationsection 3 outputs the spreading code used in a GPS satellite signal tobe received. As a result, a result of correlation between the receivedsignal and the spreading code is obtained from the digital matchedfilter 2.

This result of correlation indicates a correlation value at each chipphase in one period of the spreading code. When a spreading code in thereceived signal r(n) is synchronized with the spreading code sent fromthe spreading-code generation section 3, a correlation waveform isobtained as shown in FIG. 3, in which a correlation value at one chipphase np among 1023 chips shows a peak which exceeds a thresholddetermined in advance. The chip phase having the peak is the phase of acorrelation point. The output of the digital matched filter 2 is likethe result of correlation shown in FIG. 3, repeated every period of thespreading code.

As described before, when a correlation point is detected from oneperiod of the spreading code, the detection sensitivity is low.Therefore, in the present embodiment, the following measure is taken.

The result of correlation repeated every period of the spreading code,output from the digital matched filter 2, is sent to a unit-periodlinear addition section 4. The unit-period linear addition section 4performs linear addition such that a value at each chip phase issynchronously added in the result of correlation repeated every periodof the spreading code, output from the digital matched filter 2, withina unit period selected as a period which is a multiple of one period ofthe spreading code and which is shorter than the bit transition periodof navigation data, 20 milliseconds. The unit-period linear additionsection 4 performs this linear addition process in each unit period.

In the present embodiment, the unit period is set to 10 milliseconds,which is 10 times one period of the spreading code and which is half thebit transition period (20 milliseconds).

As shown in FIG. 2, the unit-period linear addition section 4 has thesame number of stages of a register 4RG as the number N of chips in thespreading code, and accumulates in the register 4RG the result ofcorrelation obtained at each chip phase of the spreading code, theresult being linearly added for the unit period. Since the unit periodis 10 milliseconds in this case, the stage of the register 4RG,corresponding to each chip phase linearly adds the 10 results ofcorrelation obtained at each chip phase in the unit-period linearaddition section 4.

The unit-period linear addition section 4 sends the result(corresponding to one period of the spreading code) of linear additionsof correlation values within the unit period to an absolute-valuecalculation section 5. The result is converted to an absolute valueevery unit period, and then sent to an absolute-value accumulativeaddition section 6. The absolute-value accumulative addition section 6accumulatively adds the absolute value of the result of linear additionsof correlation values for each unit period, for a period M (M is aninteger equal to two or more) times the unit period. Then, theabsolute-value accumulative addition section 6 sends the result ofaccumulative additions to a correlation-point detection section 7.

The correlation-point detection section 7 compares the result ofaccumulative additions, which characteristic is shown in FIG. 3, with athreshold determined in advance. When a peak exceeding the threshold isdetected, it is assumed that the peak means that the received signal hasbeen synchronized with the spreading code, and the phase of the peak isdetected as a correlation point np.

A reference clock is sent from a reference-clock generator 10 to ascaler 8, a clock signal CLK having the same frequency as the samplingfrequency of the received signal r(n) is generated, and the clock signalCLK is sent to the A/D converter 1, the digital matched filter 2, andthe unit-period linear addition section 4.

The reference clock is also sent from the reference-clock generator 10to a timing control section 9. The timing control section 9 generates atiming signal synchronized with the unit period, and sends it to theunit-period linear addition section 4, the absolute-value calculationsection 5, and the absolute-value accumulative addition section 6.

[Specific Example of Digital Matched Filter]

The digital matched filter 2 shown in FIG. 2 can be formed of atransversal filter or by using FFT. FIG. 4 shows an example structure ofthe digital matched filter 2 which employs a transversal filter.

The digital matched filter 2 shown in FIG. 2 has the same number ofstages of a shift register 201 as the number N of chips of the spreadingcode minus one. A digital signal Din sent from the A/D converter 1 issequentially transferred into the shift register 201 by the clock signalCLK sent from the scaler 8, not shown in FIG. 4.

The digital signal Din and the output of each register RG constitutingthe shift register 201 are multiplied by coefficients in coefficientmultipliers 202 ₁, 202 ₂, 202 ₃, . . . , and 202 _(N), and then sent toa summing unit 203 for a summing calculation. The result of the summingcalculation output from the summing unit 203 is attenuated by 1/N in alevel adjustment section 204, and output as a result CRout ofcorrelation.

The coefficient multipliers 202 ₁, 202 ₂, 202 ₃, . . . , and 202 _(N)also receive the values (+1 or −1) of chips in the spreading code fromthe spreading-code generation section 3. In this case, the values of thechips in the spreading code are sent to the coefficient multipliers in areverse order such that the first chip in the spreading code sent fromthe spreading-code generation section 3 corresponds to the coefficientmultiplier 202 _(N), and the 1023-th chip corresponds to the coefficientmultiplier 202 ₁.

Therefore, the result CRout of correlation output from the summing unit203 shows a peak at a chip phase corresponding to when the shiftregister 201 receives a digital signal synchronized with the spreadingcode sent from the spreading-code generation section 3, and the resultof correlation is in a low level at the other chip phases. In otherwords, the signal having the characteristic shown in FIG. 3 is obtainedas the result CRout of correlation output from the summing unit 203.

Next, FIG. 5 shows an example structure of the digital matched filter 2which employs FFT.

In the case shown in FIG. 5, a digital signal Din sent from the A/Dconverter 1 is written into a buffer memory 211. The signal written intothe buffer memory 211 is read in units of periods (1023 chips) of thespreading code and FFT-processed by an FFT processing section 212. Theresult of the FFT processing is written into a memory 213. The result ofthe FFT processing read from the memory 213 is sent to a multiplier 214.

The spreading-code generation section 3 generates a spreading code ofthe same series as the spreading code used in a received signal sentfrom a satellite from which signals are to be received. The spreadingcode of one period (1023 chips) sent from the spreading-code generationsection 3 is sent to an FFT processing section 215 and FFT-processed,then, the complex conjugate thereof is calculated, and the result ofprocessing is sent to a memory 216 as the result of FFT processing ofthe spreading code.

In the same way as in usual cases, the result of FFT processing issequentially read from a lower frequency from the memory 216 and sent tothe multiplier 214.

The multiplier 214 multiplies the result of FFT processing of thereceived signal sent from the memory 213 by the result of FFT processingof the spreading code sent from the memory 216 to calculate the degreeof correlation between the received signal and the spreading code in afrequency domain. The result of multiplication is sent to an inverse-FFTprocessing section 217, and the signal in the frequency domain isconverted to a signal in a time domain.

The result of inverse-FFT processing obtained from the inverse-FFTprocessing section 217 is a correlation detection signal of the receivedsignal and the spreading code in the time domain. This correlationdetection signal is sent to the unit-period linear addition section 4.

In the same way as in the above-described case, in which the digitalmatched filter employs a transversal filter, this correlation detectionsignal indicates a correlation value at each chip phase in one period ofthe spreading code. When the spreading code in the received signal issynchronized with the spreading code sent from the spreading-codegeneration section 3, a correlation waveform is obtained as shown inFIG. 3, in which a correlation value at one chip phase among 1023 chipsshows a peak which exceeds a threshold determined in advance. The chipphase having the peak is the phase of a correlation point.

The principle of the processing performed by the digital matched filterin the case shown in FIG. 5 is based on a theorem in which convolutionalFourier transform in the time domain is a multiplication in thefrequency domain, as shown in expression (1) of FIG. 22.

In expression (1), r(n) indicates a received signal in the time domain,and R(k) indicates the discrete Fourier transform thereof. In addition,c(n) indicates a spreading code sent from the spreading-code generationsection, C(K) indicates the discrete Fourier transform thereof, “n”indicates a discrete time, “k” indicates a discrete frequency, and F[ ]indicates Fourier transform.

When a correlation function of the two signals r(n) and c(n) is definedas f(n), F(k) which shows the discrete Fourier transform of f(n) has arelationship shown in expression (2) of FIG. 22. Therefore, when it isassumed that r(n) is a signal sent from the A/D converter 1 shown inFIG. 1 and c(n) is a spreading code sent from the spreading-codegeneration section 3, the correlation function f(n) of r(n) and c(n) canbe calculated in the following procedure by using expression (2) withoutusing a usual definition expression.

Calculate R(k) which is the discrete Fourier transform of the receivedsignal r(n).

Calculate the complex conjugate C(k) of C(k) which is the discreteFourier transform of the spreading code c(n).

Calculate F(k) in expression (2) by using R(k) and the complex conjugateC(k) of C(k).

Calculate a correlation function f(n) by applying inverse discreteFourier transform to F(k).

When the spreading code included in the received signal r(n) matches thespreading code c(n) sent from the spreading-code generation section 3,as described above, the correlation function f(n) calculated accordingto the above procedure has a time waveform that has a peak at acorrelation point as shown in FIG. 3. As described above, in the presentembodiment, since high-speed FFT and inverse FFT algorithms are used inthe discrete Fourier transform and the inverse Fourier transform,calculations are performed substantially faster than when correlation iscalculated according to the definition.

In the case shown in FIG. 5, the spreading-code generation section 3 andthe FFT processing section 215 are separately provided. When FFT isapplied in advance to the spreading code corresponding to each GPSsatellite, and the obtained result is stored in a memory, an FFTcalculation applied to the spreading code c(n) when a satellite signalis received can be omitted.

Description of Operations in First Embodiment

FIG. 1 is a timing chart used for describing operations in the firstembodiment having the above-described structure.

As described above, when the carrier is removed, a GPS signal isobtained by applying spectrum spreading to 50-bps navigation data with aspreading code having a frequency of 1.023 MHz and a period of 1023. Thetime length of one bit in the navigation data is 20 milliseconds, andincludes 20 periods of the spreading code which has a period of onemillisecond.

Although a noise component is much larger in the GPS signal, whencorrelation with a spreading code generated inside the receiver isdetected in many periods, the receiving sensitivity is improved. This isbecause the GPS signal is a periodic signal whereas heat noise, which isa main noise component, is a random, non-periodic signal.

The GPS signal is partially a periodic signal, but includes navigationdata which has bit transitions as shown in FIG. 1(A). Since thenavigation data is unknown, when correlation with the spreading code isobtained for many periods of the spreading code, positive and negativesignals (bit “1” and bit “0”) are offset by each other in the navigationdata, as described above, and there is a case in which a correlationpeak such as that shown in FIG. 3 is not detected.

To prevent this from occurring, as described above, there is a method inwhich the sum of the absolute values or the square values of correlationobtained every period (one millisecond) of the spreading code is addedfor many periods. According to this method, the correlation peak doesnot depend on the navigation data. However, a loss caused by squaringoperations becomes larger as C/N (electric-power ratio of carrier tonoise) becomes smaller, and the degree of improvement in the detectionsensitivity is low.

Therefore, in the first embodiment, the detection sensitivity isimproved in a way described below.

As shown in FIG. 1(B), in the first embodiment, ten periods (10milliseconds) of the spreading code, which is half the period of one bitin the navigation data, is used as a unit period. In this unit period,that is, 10 periods of the spreading code, the digital matched filter 2sequentially obtains the result of correlation calculations every period(one millisecond) of the spreading code, as described before.

Then, the unit-period linear addition section 4 linearly adds the resultof correlation calculations obtained every period of the spreading code,within the unit period.

Next, the absolute-value calculation section 5 calculates the absolutevalue of the result of linear additions of correlation values, obtainedevery unit period and sent from the unit-period linear addition section4, as shown in FIG. 1(D).

Then, the absolute-value accumulative addition section 6 adds theabsolute values for M periods (hereinafter called M zones) of thespreading code. The total sum of the absolute values in the M periods issent to the correlation-point detection section 7, and a correlationpoint is detected.

In this case, since bit transition positions in the navigation datashown in FIG. 1(A) are unknown, the unit period and the bit transitionpositions of the navigation data usually differ in phase in timing, asshown in the figure. Therefore, positive and negative components areoffset by each other in the process of linear correlation calculationswithin unit periods which include a bit transition, such as second andfourth unit periods from the left in FIG. 1(B), and especially when abit transition occurs at the center of a unit period, positive andnegative components are offset by each other completely, and the resultof correlation calculations becomes zero in the unit period.

In the present embodiment, however, since at least half a plurality ofunit periods in the M zones does not include a bit transition, adifference in the detection sensitivity is just 3 dB between the bestcase in which the navigation data has the same sign over the M zones orbit transition positions match boundaries of unit periods, and the worstcase in which the navigation data alternately has “0” and “1” in the Mzones and bit transitions are located at the centers of unit periods.

As described above, the spreading-code synchronization detection methodaccording to the present embodiment is simple, but, since a loss causedby squaring (absolute-value obtaining) operations is small, has a higherdegree of improvement in the synchronization detection sensitivity thanthe method in which the sum of the absolute values of correlationobtained in units of the periods of the spreading code is obtained in aperiod M times the unit period.

In the above-described first embodiment, the unit period, in which theresults of correlation obtained in units of periods of the spreadingcode are linearly added, is set to 10 periods of the spreading code,which is equal to half the period of one bit in the navigation data. Theunit period does not necessarily need to be set to the 10 periods. Whenthe unit period is set shorter than the 10 periods, the degree ofimprovement in the detection sensitivity is lowered but dispersion incorrelation values, caused by bit transition positions is low.Conversely, when the unit period is set longer than the 10 periods,dispersion caused by bit transition positions is high but the degree ofimprovement in the detection sensitivity becomes high depending on thebit transition positions.

Second Embodiment

A second embodiment is a modification of the first embodiment, anddiffers from the first embodiment in the procedure to obtain the resultof correlation-value linear additions.

More specifically, in the above-described first embodiment, the resultof correlation calculations is obtained every period of the spreadingcode by the digital matched filter 2, and the result of correlationcalculations is linearly added every unit period. When the receivedsignal r(n) is linearly added in one period of the spreading code beforethe digital matched filter 2 performs correlation calculations, and thedigital matched filter applies correlation calculations to the result oflinear additions, the exactly same result and advantage as in theabove-described case are obtained. This is a case described in thesecond embodiment.

FIG. 6 is a block diagram showing an example structure of aspreading-code synchronization detection apparatus according to thesecond embodiment.

In the second embodiment, a digital signal output from an A/D converter1 is sent to a unit-period linear addition section 11. The unit-periodlinear addition section 11 linearly adds a digital signal obtained in aunit period, actually in this case, a digital signal obtained in 10periods (10 milliseconds) of the spreading code, in one period of thespreading code. In other words, 10 data items at the same chip phase ofthe spreading code are synchronously added in the digital signal in 10periods of the spreading code, within each unit period.

Therefore, the unit-period linear addition section 11 outputs the samenumber of the results (the same number of data items as the number ofchips in one period of the spreading code) of synchronous additions asthe number of data items in one period of the spreading code. Theresults of synchronous additions are sent to the digital matched filter2. Correlation calculations are applied to the results of synchronousadditions and the spreading code sent from a spreading-code generationsection 3. The result of correlation calculations is sent to anabsolute-value calculation section 5. The other structure is the same asthat of the first embodiment.

The second embodiment differs from the first embodiment in that linearadditions are performed every period of the spreading code in a stageprior to the digital matched filter 2, but obtains exactly the sameresult and advantage as the first embodiment.

Also in the second embodiment, the unit period is set to 10 periods ofthe spreading code, which is half the one-bit period of navigation data.The unit period is not necessarily equal to the 10 periods, as in theabove-described first embodiment.

Third Embodiment

A third embodiment is also a modification of the first embodiment, anddiffers from the first embodiment in the procedure to obtain the resultof correlation-value linear additions.

In the third embodiment, in a case in which a digital matched filterusing FFT, as shown in FIG. 5, is used as the digital matched filter 2,the unit-period linear addition section 4 disposed subsequent to thedigital matched filter 2 in the first embodiment, or the unit-periodlinear addition section 11 disposed before the digital matched filter 2in the second embodiment is omitted.

FIG. 7 is a block diagram showing an example structure of aspreading-code synchronization detection apparatus according to thethird embodiment.

More specifically, in the third embodiment, a digital signal output froman A/D converter 1 is sent to a digital matched filter 12 using FFT,formed of a portion enclosed by a dotted line in FIG. 5.

In the above-described first embodiment, a digital signal is read everyperiod of the spreading code from the memory 211 and sent to the FFTprocessing section 212, as described by referring to FIG. 5. In thethird embodiment, digital data is read from a memory 211 every unitperiod, and sent to an FFT processing section 212, and FFT calculationsare performed in the digital matched filter 12 using FFT.

In the third embodiment, the FFT processing section 212 applies FFTcalculations to a digital signal obtained every unit period. Since theunit period includes 10 periods of the spreading code in theabove-described case, the FFT processing section 212 outputs the resultof FFT calculations, which is the same as the result obtained byaccumulating the result of FFT calculations of the digital signalobtained every period of the spreading code, for 10 periods, and theoutput result of FFT calculations is written into a memory 213.

Subsequent processes in the digital matched filter 12 using FFT areexactly the same as those described by referring to FIG. 5. An inverseFFT processing section 217 outputs the result of correlationcalculations, which is again expressed in the time domain. In the thirdembodiment, the result of correlation calculations is output from thedigital matched filter 12 using FFT to an absolute-value calculationsection 5.

As described above, according to the third embodiment, since FFTcalculations are applied to a digital signal obtained in a unit period,the unit-period linear addition section 4 disposed subsequent to thedigital matched filter 2 in FIG. 2, or the unit-period linear additionsection 11 disposed before the digital matched filter 2 in FIG. 6 can beomitted, and the structure is simplified.

Even in the second embodiment, the unit period is set to 10 periods ofthe spreading code, which is half a one-bit period of the navigationdata. Exactly in the same way as in the above-described firstembodiment, the unit period does not necessarily include the 10 periods.

Fourth Embodiment

In the above-described first to third embodiments, the method is used,in which the results of correlation calculations obtained every periodof the spreading code are linearly added, the absolute values of theresults of linear additions are accumulatively added in the M zones, anda correlation point is detected from the result of accumulativeadditions. Therefore, the higher degree of improvement in the detectionsensitivity of a correlation point is obtained than in the conventionalmethod described at the top part of the specification.

In the method used in the above-described first to third embodiments,however, since the results of correlation calculations may havedispersion depending on the phase relationship between the unit periodand bit transition positions, it is difficult to set a threshold withrespect to a peak correlation value when a correlation point isdetected. Therefore, it is difficult to use the method in terms of thedetermination of whether correlation exists. A fourth embodiment solvesthis problem.

FIG. 8 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to the fourthembodiment. FIG. 9 is a timing chart used for describing the operationof the apparatus shown in FIG. 8.

In the fourth embodiment, two signal series are generated from areceived signal r(n) (shown in FIG. 9(A)). A first signal series A(shown in FIG. 9(B)) is the same as the received signal r(n).

As for the first signal series A, a digital signal output from an A/Dconverter 1 is sent as is to a digital matched filter 21, andcorrelation calculations with a spreading code sent from aspreading-code generation section 3 are performed, as described in theabove-described first embodiment. The result of correlation calculationsfor each period of the spreading code is sent to a unit-period linearaddition section 22, and linearly added in a unit period.

In the fourth embodiment, the unit period is set to include one bit ofthe navigation data, that is, 20 periods of the spreading code.

A linear-addition correlation-calculation result DA (shown in FIG. 9(C))output from the unit-period linear addition section 22 is sent to anabsolute-value calculation section 23, and is converted to its absolutevalue, and is sent to an addition section 24. The addition output D ofthe addition section 24 is sent to an accumulative addition section 25,and accumulatively added over M zones in the same way as in the firstembodiment. The result MD of accumulative additions is sent to acorrelation-point detection section 26.

A clock signal CLK output from an scaler 8 and various timing signalsoutput from a timing control section 9 are sent to circuit blocks in thesame way as in the above-described first embodiment. As described above,the fourth embodiment differs from the first embodiment in that the unitperiod is 20 milliseconds.

The structure used for the first signal series A is the same as thatused in the above-described first embodiment except the addition section24. The difference is that the unit period includes the period of onebit of the navigation data, that is, 20 milliseconds in the fourthembodiment, whereas the unit period is half the period of one bit of thenavigation data, that is, 10 milliseconds in the above-described firstembodiment.

The result of linear additions for 10 milliseconds can be added twice toobtain the result of linear additions for 20 milliseconds for the firstsignal series A, in the same way as in the first embodiment.

Next, a process for a second signal series B will be described. As forthe second signal series B, the digital signal output from the A/Dconverter 1 is sent to one input end of a switch circuit 35 as is, andalso sent to a sign inversion section 34, its sign is inverted, and thensent to the other input end of the switch circuit 35.

The switch circuit 35 is alternately switched by a switch switchingsignal SW sent from the timing control section 9 between the one inputend for the first half period (10 milliseconds) of a unit period and theother input end for the second half period (10 milliseconds) of the unitperiod. Therefore, the switch circuit 35 outputs the signal series B(shown in FIG. 9(D)) in which the digital signal converted from thereceived signal r(n) is inverted in sign at the second half period of aunit period.

The signal series B digital signal is sent to a digital matched filter31, and correlation calculations with the spreading code sent from thespreading-code generation section 3 are performed, as described in theabove-described first embodiment. The result of correlation calculationsperformed every period of the spreading code is sent to a unit-periodlinear addition section 32, and linearly added in a unit period.

A linear-addition correlation-calculation result DB (shown in FIG. 9(E))output from the unit-period linear addition section 32 is sent to anabsolute-value calculation section 33, is converted to its absolutevalue, and then sent to the addition section 24. The absolute value ofthe result DB is added to the absolute value of the linear-additioncorrelation-calculations result DA, output from the absolute-valuecalculation section 23. Therefore, the result D of additions output fromthe addition section 24 is:D=|DA|+|DB|  [expression (3)]

The result D of additions is accumulated for the M zones (M.≧1) by anaccumulative addition section 25, and the result MD of accumulation issent to a correlation-point detection section 26. The correlation-pointdetection section 26 determines whether a peak larger than a thresholdspecified in advance has been detected to determine whether acorrelation point has been detected. When a peak is detected, the chipphase where the peak is obtained is detected as a correlation point np.

When the phase shift between a bit transition position and a unit periodis called “h” as shown in FIG. 9(A) in the fourth embodiment, thelinear-addition correlation-calculation results DA and DB output fromthe unit-period linear addition sections 22 and 32 with respect to thephase shift “h” are shown in FIG. 10A and FIG. 10B.

FIG. 10A and FIG. 10B are based on a case in which the navigation datais inverted every bit. In addition, FIG. 10A and FIG. 10B arecharacteristic views assuming that a correlation value obtained in oneperiod of the spreading code is “d” and “M” is set to one. FIG. 10Ashows a case in which the correlation value “d” is negative, and FIG.10B shows a case in which the correlation value “d” is positive. Thephase shift “h” is expressed in the number of periods of the spreadingcode (can be expressed in milliseconds because one period of thespreading code is one millisecond).

More specifically, as shown in FIG. 10A and FIG. 10B, when the topposition of a unit period is synchronized with a bit transition positionof the navigation data at a phase shift “h” of zero, since 20correlation values “d” each obtained in one period of the spreadingcode, in a unit period are all positive or all negative in the firstsignal series A, the linear-addition correlation-calculation result DAis:DA=20|d| when d>0DA=−20|d| when d<0

In contrast, since the sign is inverted at the center of a unit periodin the second signal series B when the phase shift “h” is zero, 20correlation values “d” each obtained in one period of the spreadingcode, in a unit period are half positive and half negative and offset byeach other. Therefore, the linear-addition correlation-calculationresult DB becomes zero.

When a bit transition position is at the center of a unit period at aphase shift “h” of 10, since 20 correlation values “d” each obtained inone period of the spreading code, in a unit period are half positive andhalf negative by bit inversion in the first signal series A, and areoffset by each other. Therefore, the linear-additioncorrelation-calculation result DA becomes zero.

In contrast, since the sign is inverted synchronously with a bittransition at the center of a unit period when the phase shift “h” is 10in the second signal series B, 20 correlation values “d” each obtainedin one period of the spreading code, in a unit period are all positiveor all negative. The linear-addition correlation-calculation result DBis:DB=−20|d| when d>0DB=20|d| when d<0

When the unit period is shifted by one bit of the navigation data at aphase shift “h” of 20, since 20 correlation values “d” each obtained inone period of the spreading code, in a unit period are all positive orall negative in the first signal series A, although the polarity isinverted from when the phase shift “h” is zero, the linear-additioncorrelation-calculation result DA is:DA=−20|D| when d>0DA=20|D| when d<0

In contrast, since the sign is inverted at the center of a unit periodin the second signal series B when the phase shift “h” is 20, 20correlation values “d” each obtained in one period of the spreadingcode, in a unit period are half positive and half negative, although thepolarity is inverted from when the phase shift “h” is zero, and thecorrelation values are offset by each other. Therefore, thelinear-addition correlation-calculation result DB becomes zero.

Since the linear-addition correlation-calculation result DA and thelinear-addition correlation-calculation result DB show thecharacteristics shown in FIG. 10A and FIG. 10B with respect to the phaseshift “h,” which is the phase difference of the bit transition positionof the navigation data with respect to the top position of a unitperiod, the sum D of the absolute values of both the linear-additioncorrelation-calculation result DA and the linear-additioncorrelation-calculation result DB becomes constant. In other words, inthis case, the following expression is satisfied.D=|DA|+|DB|=20|d|

In other words, the sum D of the absolute values of both thelinear-addition correlation-calculation result DA and thelinear-addition correlation-calculation result DB is constant, 20M|d|,irrespective of the phase shift “h” between the bit transition positionsof the received signal r(n) and the unit period. Therefore, the resultMD of accumulation, output from the accumulative addition section 25 isa value obtained by simply accumulatively adding the constant for the Mzones, that is:MD=|DA|+|DB|=20M|d|

As described above, according to the fourth embodiment, even when thebit transition position is disposed anywhere with respect to the topposition of a unit period, since the sum of the absolute values becomesa multiple of the correlation value “d,” it becomes easier to specifythe threshold in the correlation-point detection section 26, used fordetermining whether correlation exists. In addition, since DA and DB arethe results of linear additions of correlation values each obtained inone period of the spreading code, in a unit period, the correlationvalues are added while noise is removed. It is expected that thedetection sensitivity of a correlation point is improved.

In the fourth embodiment, as the digital matched filters 21 and 31, atransversal filter like that shown in FIG. 4 or a structure using an FFTprocess like that shown in FIG. 5 can be used in the same way as in theabove-described first to third embodiments.

In the above description of the fourth embodiment, the unit period isset equal to the bit transition period. In the fourth embodiment, theunit period needs to be the bit transition period or shorter.

Fifth Embodiment

A fifth embodiment is a modification of the fourth embodiment and hasthe same relationship with the fourth embodiment as that of the secondembodiment with the first embodiment. The fifth embodiment differs fromthe fourth embodiment in the procedure to obtain the result ofcorrelation-value linear additions.

More specifically, in the above-described fourth embodiment, correlationcalculations are performed every period of the spreading code for thefirst signal series A and the second signal series B by the digitalmatched filters 21 and 31, and the results of correlation calculationsare linearly added every unit period. In the fifth embodiment, a firstsignal series A and a second signal series B are respectively linearlyadded in one period of the spreading code before the digital matchedfilters 21 and 31 perform correlation calculations, and the digitalmatched filters 21 and 31 apply correlation calculations to the resultsof linear additions.

FIG. 11 is a block diagram showing an example structure of aspreading-code synchronization detection apparatus according to thefifth embodiment.

In the fifth embodiment, the first series A digital signal output froman A/D converter 1 is sent to a unit-period linear addition section 27.The unit-period linear addition section 27 applies linear additions inone period of the spreading code to a digital signal obtained in a unitperiod, actually in this case, a digital signal obtained in 20 periods(20 milliseconds) of the spreading code. In other words, 20 data itemsat the same chip phase of the spreading code are synchronously added inthe digital signal in 20 periods of the spreading code within each unitperiod.

The second series B digital signal output from a switch circuit 35 issent to a unit-period linear addition section 36. Like the unit-periodlinear addition section 27, the unit-period linear addition section 36applies linear additions in one period of the spreading code to adigital signal obtained in a unit period, actually in this case, adigital signal obtained in 20 periods (20 milliseconds) of the spreadingcode. In other words, 20 data items at the same chip phase of thespreading code are synchronously added in the digital signal in 20periods of the spreading code within each unit period.

Then, the results of synchronous additions (the same number of dataitems as the number of chips in one period of the spreading code) aresent to the digital matched filters 21 and 31, and correlationcalculations are applied to the results of synchronous additions and thespreading code sent from a spreading-code generation section 3. Theresults of correlation calculations are sent to an absolute-valuecalculation sections 23 and 33. The other structure is the same as thatof the fourth embodiment.

The fifth embodiment differs from the fourth embodiment in that linearadditions are performed in one period of the spreading code in a stageprior to the digital matched filters 21 and 31, but obtains exactly thesame result and advantage as the fourth embodiment.

Also in the fifth embodiment, the unit period is not limited to a casein which the unit period is set equal to the bit transition period. Theunit period needs to be set equal to the bit transition period orshorter.

Sixth Embodiment

A sixth embodiment is a modification of the fourth embodiment and hasthe same relationship with the fourth embodiment as that of the thirdembodiment with the first embodiment. The sixth embodiment differs fromthe fourth embodiment in the procedure to obtain the result ofcorrelation-value linear additions.

In the sixth embodiment, in a case in which a digital matched filterusing FFT like that shown in FIG. 5 is used as digital matched filter 21and 31, the unit-period linear addition sections 22 and 32 disposedsubsequent to the digital matched filters 21 and 31 in the fourthembodiment, or the unit-period linear addition sections 27 and 36disposed before the digital matched filters 21 and 31 in the fifthembodiment are omitted.

FIG. 12 is a block diagram showing an example structure of aspreading-code synchronization detection apparatus according to thesixth embodiment.

More specifically, in the sixth embodiment, a first signal series Adigital signal is sent to a digital matched filter 28 using FFT, formedof a portion enclosed by a dotted line in FIG. 5, and written into itsmemory 211. In the same way, a second signal series B digital signal issent to a digital matched filter 37 using FFT, and written into itsmemory 211.

In the sixth embodiment, digital data is read from the memory 211 in aunit period, and the digital data read in the unit period is sent to anFFT processing section 212, and FFT calculations are performed in eachof the digital matched filters 28 and 37 using FFT.

In the sixth embodiment, the FFT processing sections 212 of the digitalmatched filters 28 and 37 apply FFT calculations to digital signalsobtained for a unit period in the signal series A and the signal seriesB, respectively. Since the unit period includes the signal series A andthe signal series B corresponding to 20 periods of the spreading code inthe above-described case, each FFT processing section 212 outputs theresult of FFT calculations, which is the same as the result obtained byaccumulating the result of FFT calculations of the digital signalobtained every period of the spreading code, for 10 periods, and theoutput result of FFT calculations is written into a memory 213.

The subsequent processes in the digital matched filters 28 and 36 usingFFT are exactly the same as those described by referring to FIG. 5. Aninverse FFT processing section 217 outputs the result of a correlationcalculation, which is again expressed in the time domain. In the sixthembodiment, the results of correlation calculations output from thedigital matched filters 28 and 36 using FFT are the linear-additioncorrelation-calculation results DA and DB, and are sent toabsolute-value calculation sections 23 and 33. The other structure isthe same as that in the fourth embodiment.

As described above, according to the sixth embodiment, since FFTcalculations are applied to digital signals obtained in a unit period,the unit-period linear addition sections 22 and 32 disposed subsequentto the digital matched filters 21 and 31 in FIG. 8, or the unit-periodlinear addition sections 27 and 36 disposed before the digital matchedfilters 21 and 31 in FIG. 11 can be omitted, and the structure issimplified.

Also in the sixth embodiment, the unit period is not limited to a casein which the unit period is set equal to the bit transition period.Since half a unit period needs to be equal to the bit transition periodor shorter, the unit period needs to be twice the bit transition periodor shorter.

Seventh Embodiment

In the above-described fourth to sixth embodiments, a digital signalconverted from the received signal r(n) is inserted in sign between thefirst half and the second half of a unit period to generate the secondsignal series B, and the linear-addition correlation-calculation resultDB is obtained in a unit period. Instead of inverting in sign thedigital signal converted from the received signal r(n) between the firsthalf and the second half of a unit period, when the spreading code sentfrom the spreading-code generation section 3 is inverted in sign betweenthe first half and the second half of a unit period and sent to adigital matched filter, the same result and advantage are obtained.

The seventh embodiment shows this case. FIG. 13 is a block diagramshowing an example structure of a spreading-code synchronizationdetection apparatus according to the seventh embodiment.

More specifically, a digital signal output from an A/D converter 1 issent to a digital matched filter 41A for a first signal series A. Thedigital matched filter 41A receives the spreading code as is from thespreading-code generation section 3. Therefore, the digital matchedfilter 41A outputs the same results of correlation calculations as thedigital matched filter 21 for the first signal series A in the fourthembodiment, shown in FIG. 8.

The results of correlation calculations, each obtained in the period ofthe spreading code are sent from the digital matched filter 41A to aunit-period linear addition section 22, and linearly added for a unitperiod, in this case, for a one-bit period (20 milliseconds) of thenavigation data in the same way as in the fourth embodiment. Thelinear-addition correlation-calculation result DA is sent from theunit-period linear addition section 22 to an absolute-value calculationsection 23, converted to its absolute value, and sent to an additionsection 24.

The digital signal output from the A/D converter 1 is also sent to adigital matched filter 41B for a second signal series B. The digitalmatched filter 41B receives the spreading code which is inverted in signbetween the first half and the second half of a unit period, from aswitch circuit 43.

More specifically, the spreading code sent from the spreading-codegeneration section 3 is input to one input end of the switch circuit 43as is, and is also input to a sign inversion section 42, inverted insign, and sent to the other input end of the switch circuit 43. Theswitch circuit 43 is alternately switched by a switching signal SW sentfrom a timing control section 9 between the one input end for the firsthalf of a unit period and the other input end for the second half of theunit period.

As described above, since the spreading code sent to the digital matchedfilter 41B is inverted in sign between the first half and the secondhalf of a unit period, the digital matched filter outputs exactly thesame results of correlation calculations as the digital matched filter31 for the second signal series B in the fourth embodiment, shown inFIG. 8.

The results of correlation calculations, each obtained in the period ofthe spreading code are sent from the digital matched filter 41A to theunit-period linear addition section 22, and linearly added for a unitperiod, in this case, for a one-bit period (20 milliseconds) of thenavigation data in the same way as in the fourth embodiment. Thelinear-addition correlation-calculation result DA is sent from theunit-period linear addition section 22 to the absolute-value calculationsection 23, converted to its absolute value, and sent to the additionsection 24.

In the same way, the results of correlation calculations, each obtainedin the period of the spreading code are sent from the digital matchedfilter 41B to a unit-period linear addition section 32, and linearlyadded for a unit period, in this case, for a one-bit period (20milliseconds) of the navigation data in the same way as in the fourthembodiment. The linear-addition correlation-calculation result DB aresent from the unit-period linear addition section 32 to anabsolute-value calculation section 33, converted to its absolute value,and sent to the addition section 24.

Therefore, the addition section 24 outputs the sum D of the absolutevalues, D=|DA|+|DB|. An accumulative addition section 25 calculates theresult MD of accumulative additions thereof for M zones, and acorrelation-point detection section 26 detects a correlation point forthe result MD of accumulative additions. In other words, the same resultand advantage as those in the fourth embodiment are obtained.

The method used in the seventh embodiment, in which the spreading codeis inverted in sign between the first half and the second half of a unitperiod to obtain the results of correlation calculations of the secondsignal series B, can also be applied to the fifth embodiment.

More specifically, although not shown in a figure, one half-unit-periodlinear addition section is provided at the output side of the A/Dconverter 1 in FIG. 13. The half-unit-period linear addition sectionlinearly adds data in the digital signal, corresponding to each chip inthe spreading code for the first half and the second half of a unitperiod, that is, for half a unit period, in synchronization with theswitching performed by the switch circuit 43 to linearly add the digitalsignal. The results of linear additions are sent to the digital matchedfilters 41A and 41B.

In this case, the digital matched filters 41A and 41B output the sameresults of correlation calculations as the linear-additioncorrelation-calculation results DA and DB, described above, andtherefore, the unit-period linear addition sections 22 and 32 areunnecessary. The apparatus is structured such that the digital matchedfilters 41A and 41B output the results of correlation calculations tothe absolute-value calculation sections 23 and 33 shown in FIG. 13.

The method used in the seventh embodiment, in which the spreading codeis inverted in sign between the first half and the second half of a unitperiod to obtain the results of correlation calculations of the secondsignal series B, can further be applied to the sixth embodiment.

More specifically, although not shown in a figure also in this case, thedigital matched filters 41A and 41B shown in FIG. 13 are formed ofdigital matched filters using FFT. The digital matched filters using FFTperform correlation calculations between the digital signal and thespreading code sent from the spreading-code generation section 3 or fromthe switch circuit 43 with the digital signal obtained in a unit periodbeing used as an FFT calculation unit, as described by referring to FIG.5 and FIG. 12.

According to this case, since FFT calculations are applied to thedigital signal obtained in a unit period in the same way as in the caseshown in FIG. 11, the unit-period linear addition sections disposedbefore or after the digital matched filters can be omitted, and thestructure is simplified.

Also in the seventh embodiment, the unit period is not limited to a casein which the unit period is set equal to the bit transition period.Since half the unit period needs to be the bit transition period orshorter, the unit period needs to be twice the bit transition period orshorter.

Eighth Embodiment

An eighth embodiment shows another case in which the linear-additioncorrelation-calculation results DA and DB are obtained. In the fourth toseventh embodiments, digital matched filters for two paths are required.In the eighth embodiment, a structure which requires just one digitalmatched filter is proposed.

FIG. 14 is a block diagram of a spreading-code synchronization detectionapparatus for a spectrum spreading signal according to the eighthembodiment. FIG. 15 is a timing chart used for describing operationsaccording to the eighth embodiment.

In the eighth embodiment, a structure from the input of a receivedsignal r(n) to a digital matched filter 2 is exactly the same as in thefirst embodiment. The digital matched filter 2 outputs the results ofcorrelation calculations each obtained in one period of the spreadingcode, to a half-unit-period linear addition section 51. Also in theeighth embodiment, a unit period is set to the bit transition period ofnavigation data, which is 20 milliseconds, in the same way as in thefourth to seventh embodiments, described above.

The half-unit-period linear addition section 51 linearly adds theresults of correlation calculations sent from the digital matched filter2 for half the unit period, that is, for a first-half period and asecond-half period of the unit period, and sends the result of linearadditions to a linear-addition-result addition section 52 and to alinear-addition-result subtraction section 53.

The linear-addition-result addition section 52 adds the result of linearadditions of the correlation values each obtained in one period of thespreading code, in the first half of a unit period to the result oflinear additions of the correlation values each obtained in one periodof the spreading code, in the second half of the unit period to generatethe linear-addition correlation-calculation result DA for the firstsignal series A, described above, as shown in FIG. 15(B) and FIG. 15(C).

The linear-addition-result subtraction section 53 subtracts the resultof linear additions of the correlation values each obtained in oneperiod of the spreading code, in the second half of the unit period fromthe result of linear additions of the correlation values each obtainedin one period of the spreading code, in the first half of the unitperiod to generate the linear-addition correlation-calculation result DBfor the second signal series B, described above, as shown in FIG. 15(D)and FIG. 15(E).

The linear-addition correlation-calculation result DA output from thelinear-addition-result addition section 52 is converted to its absolutevalue in an absolute-value calculation section 54, and then sent to anaddition section 56. The linear-addition correlation-calculation resultDB output from the linear-addition-result subtraction section 53 isconverted to its absolute value in an absolute-value calculation section55, and then sent to the addition section 56. Therefore, the sum D ofthe absolute values of the linear-addition correlation-calculationresults DA and DB for the first and second series A and B is obtainedfrom the addition section 56.

The sum D of the absolute values is sent from the addition section 56 toan accumulative addition section 57, and accumulatively added in Mzones. The result MD of accumulative additions is sent to acorrelation-point detection section 58. When a peak exceeding athreshold value specified in advance is detected, a correlation point npis detected.

According to the eighth embodiment, since the digital matched filter andthe linear addition section for linearly adding the results ofcorrelation calculations each obtained in one period of the spreadingcode can be shared for the first series A and the second series B, thestructure of the apparatus is made simple.

Also in the eighth embodiment, the unit period is not limited to a casein which the unit period is set equal to the bit transition period. Theunit period needs to be the bit transition period or shorter.

Ninth Embodiment

As described before, the linear-addition correlation-calculation resultsDA and DB for the first and second series A and B, obtained in thefourth to eighth embodiments have the characteristics shown in FIG. 10Aand FIG. 10B, described before, with respect to the phase shift “h”between a unit period and bit transitions positions of navigation data,and the sum of the absolute values of the linear-additioncorrelation-calculation results DA and DB is constant, as describedbefore.

However, the detection sensitivity of spreading-code synchronization isnot constant in the viewpoint of C/N. When a bit transition position islocated at the center of a first half or a second half of a unit period,since correlation values are offset by each other and become zero in thehalf of the unit period, where the bit transition is located,correlation values are substantially obtained only in the half of theunit period, where the bit transition is not located. Therefore, the C/Nis reduced by 3 dB and the detection sensitivity is also reducedcompared with a case in which the bit transition position matches thetop position of the unit period.

To improve this problem, the phase shift “h” needs to be compensated forsuch that the bit transition position matches the top position of theunit period.

It is understood from the characteristic views of FIG. 10A and FIG. 10Bthat the ratio DA/DB between the linear-addition correlation-calculationresults DA and DB has a unique value at each position of the phase shift“h.” Therefore, the phase shift “h” of the bit transition position fromthe top position of the unit period can be estimated from the followingexpressions.h=L/2×[1+1/{(DB/DA)−1}]  [Expression (4-1)]when DA/DB≦0 (h≦L/2)h=L/2×[1+1/{(DB/DA)+1}]  [Expression (4-2)]when DA/DB>0 (h≧L/2)Where L indicates the length of the unit period.

In the ninth embodiment, the top position of the unit period issynchronized with the bit transition position to improve the detectionsensitivity with the use of the above-described point. FIG. 16 is ablock diagram of a spreading-code synchronization detection apparatusfor a spectrum spreading code according to the ninth embodiment. Theabove-described point is applied to the eighth embodiment, describedbefore.

More specifically, in the ninth embodiment, the linear-additioncorrelation-calculation result DA output from a linear-addition-resultaddition section 52 and the linear-addition correlation-calculationresult DB output from a linear-addition-result subtraction section 53are sent to a phase-shift “h” estimation section 59. A timing controlsection 9 sends the same timing signal to the phase-shift “h” estimationsection 59 as that sent to an accumulative addition section 57.

The phase-shift “h” estimation section 59 uses the above-describedexpressions (4-1) and (4-2) to estimate the phase shift “h” of a bittransition position from the top position of each unit period in eachunit period of M zones. The M phase-shift “h” estimation values eachestimated by the phase-shift “h” estimation section 59 for each unitperiod have two distributions, as shown in FIG. 17, one centered at h=0,which shows no phase shift “h,” and the other centered at the phaseshift “h” corresponding to the actual bit transition position shiftedfrom the top position of a unit period.

The phase-shift “h” estimation section 59 determines groups related tothe two phase-shift “h” distributions, estimates the phase shift “h” ofthe bit transition position from the top position of a unit period, andcontrols the timing control section 9 according to the result ofestimation such that the timing at which the received signal r(n) istaken is shifted by the estimation value in the next and subsequentcorrelation detection so as to match the bit transition position withthe top position of the above-described unit period used for correlationdetection.

When the bit transition position is compensated as described above, theabove-described unit period used for correlation detection issynchronized with the navigation data, and a reduction in C/N can beavoided thereafter.

When C/N is at a level sufficient for demodulating the navigation data,the bit transition position which has been compensated for the phaseshift “h” matches the top position of the unit period, or slightlyshifted therefrom. Therefore, DB is distributed at zero and in thevicinity thereof, and DA is distributed at +20|d| and in the vicinitythereof, or −20|d| and in the vicinity thereof depending on whether thenavigation data bit is “0” or “1.” Consequently, whether the navigationdata bit is “0” or “1” can be determined by whether DA is +20|d| or−20|d|, that is, DA has a positive sign or a negative sign.

FIG. 18 is a flowchart for describing the flow of the above-describedprocess according to the ninth embodiment. In the process shown in theflowchart of FIG. 18, first and second signal series A and B aregenerated from a received signal, the linear-additioncorrelation-calculation results DA and DB are obtained, and the sum D ofthe absolute values thereof is further obtained as in theabove-described fourth to sixth embodiments. In this case, it is assumedthat the spreading signal is synchronized with the carrier.

This flowchart also shows a processing procedure for a case in which aspreading-code synchronization detection process is executed by softwareby using, for example, a DSP (digital signal processor) or amicrocomputer.

First, a variable “m” which indicates that the processing is applied tothe m-th unit period among M unit periods from the 0-th period to the(M−1)-th period in M zones is set to an initial value, zero, and avariable “k” which indicates the k-th phase shift “h” is set to aninitial value, zero (in step S1).

Then, a received signal r(n) converted to a digital signal in an A/Dconverter 1 is used as an input signal, and it is read into a memory forM zones (in step S2). Next, a first signal series A and a second signalseries B are generated from the digital signal obtained in a first unitperiod (m=0) (in step S3). Then, the linear-additioncorrelation-calculation results DA and DB are obtained for the series Aand B, and further the sum D of the absolute values thereof is obtained,as described before (in step S4).

Then, it is determined whether the obtained sum D of the absolute valuesin the unit period is larger than a threshold Dth specified in advance(in step S5). When it is determined that the sum D of the absolutevalues is not larger than the threshold Dth, it is determined thatcarrier synchronization is not obtained, and the procedure proceeds to aroutine for recapturing carrier synchronization.

When it is determined that the sum D of the absolute values is largerthan the threshold Dth specified in advance, the ratio DB/DA between thecorrelation-value linear-addition correlation-calculation results DA andDB is obtained, and a phase shift “h” is calculated by theabove-described expressions (4-1) and (4-2) (in step S6). Then, it isdetermined whether the sign of the linear-additioncorrelation-calculation result DA is positive or negative (in step S7).The result of determination is sent to a routine for determining andprocessing bit data of a navigation message.

Next, it is determined whether the phase shift “h” obtained in step S6is in a vicinity of zero (in step S8). The method of determination is,as described in FIG. 18, whether one of the following expressions issatisfied.h<ε  [Expression (5)]h>20−ε  [Expression (6)]

Where, ε is a value which can be regarded as a vicinity of zero, such as1.0 (millisecond), that is, is set to about one period of the spreadingcode. Since the unit period is 20 times the period of the spreadingcode, 20 milliseconds, “20−ε” in expression (6) indicates that the phaseshift is in a vicinity of one bit, that is, that the phase shift “h”between the unit period and the bit transition position is zero.

When it is determined in step S8 that the phase shift “h” is not in avicinity of zero, it is determined that the top position of the unitperiod has the phase shift “h” with respect to the bit transitionposition, the phase shift “h” is stored in a memory as “h_(k)”, and thevariable “k” is incremented by one (in step S9). Then, the procedureproceeds to the next step S10, and it is determined whether theabove-described processes have been performed for all unit periods inthe M zones.

When it is determined in step S8 that the phase shift “h” is in avicinity of zero, it is determined that the top position of the unitperiod is synchronized with the bit transition position, the procedureskips step 9 and jumps to step S10, and it is determined whether theabove-described processes have been performed for all unit periods inthe M zones.

When it is determined in step S10 that the above-described processeshave not been performed for all unit periods in the M zones, thevariable “m,” which indicates the number of a unit period, isincremented by one (in step S11) to specify the next unit period, theprocedure returns to step S3, and the processes of step S3 andsubsequent steps are repeated.

When it is determined in step S10 that the above-described processeshave been performed for all unit periods in the M zones, the phase shift“h” is estimated from the distribution state of k phase shifts h.sub.0to h.sub.k−1, by using the relationship shown in FIG. 17 (in step S12).

Sampling points of the digital signal taken in the unit period arecompensated by using the estimated phase shift “h” (in step S13), sothat the top position of the unit period is synchronized with the bittransition position. Then, the procedure returns to step S1, and theabove-described processes are repeated for the next M zones.

As described above, in the ninth embodiment, since the top position of aunit period is synchronized with a bit transition position of thenavigation data, not only the detection sensitivity of thespreading-code synchronization is improved but also the demodulation ofthe navigation data is allowed by the sign, positive or negative, of thelinear-addition correlation-calculation result DA.

In the example processing shown in the flowchart of FIG. 18, thereceived signal is stored for the M zones and then processed. Thereceived signal is not necessarily stored, but may be processed in unitsof unit zones.

Also in the ninth embodiment, the same point as that for theabove-described fourth to eighth embodiments is applied to the length ofthe unit period.

Tenth Embodiment

According to the method used in the above embodiment, the detectionsensitivity (corresponding to the receiving sensitivity of a GPSreceiver) of a spreading-code synchronization for a spectrum spreadingsignal is improved. The detection sensitivity and a processing time aretrade-offs. When correlation is detected for many periods of thespreading code to improve the detection sensitivity, the processing timeis inevitably extended.

In the above-described embodiment, it is assumed that carriersynchronization has been obtain beforehand. When a carrier frequency isunknown, a process for synchronizing the carrier is required, and anoperation for searching for the carrier frequency is provided in someform. Since a correlation value is calculated for each frequency used inthe searching, if the searching is performed many times, the GPSreceiver may show a slow response.

A tenth embodiment satisfies the improvement of the detectionsensitivity and high-speed processing at the same time. FIG. 19 is ablock diagram of a spreading-code synchronization detection apparatusfor a spectrum spreading signal according to the tenth embodiment. Whenthe tenth embodiment is applied to the apparatus of the firstembodiment, shown in FIG. 2, the case shown in FIG. 19 is obtained.

In the tenth embodiment, as shown in FIG. 19, a digital signal convertedfrom a received signal r(n) and output from an A/D converter 1 is storedin a memory 71 in, for example, M zones. A clock signal CLK having thefrequency corresponding to a chip rate of 1.023 MHz of a GPS signal andoutput from a scaler 8 is used as a writing clock signal, as in theabove-described embodiments.

In the tenth embodiment, a higher-speed clock signal CLKa having ahigher transfer speed than the chip rate of 1.023 MHz of the GPS signalis used for reading the signal from the memory 61 and for sending it toa subsequence digital matched filter 2. More specifically, a referenceclock signal is output from a reference-clock generator 10 to a scaler76, and the scaler 76 generates the higher-speed clock signal CLKa, andsends it to the memory 71 and to the digital matched filter 2.

Data read from the memory 71 is sent to a multiplier 72 for carriersynchronization. Clock signals I and Q having different phases areoutput from a clock generator 74 formed of a numerical-controlvariable-frequency oscillator (hereinafter called an NCO) to an I/Qselection section 75, alternately selected in a time-division mannertherein, and sent to the multiplier 72. The clock generator 74 receivesthe reference clock signal from the reference-clock generator 10.

A carrier control section 73 is also provided. A control signal outputfrom the carrier control section 73 controls the oscillating frequencyof the NCO 74. The carrier control section 73 is controlled by a controlsignal corresponding to the result of correlation-point detection outputfrom a correlation-point detection section 7, as described later.

While carrier synchronization is being obtained, the multiplier 72outputs a digital signal from which the carrier component has beenremoved, to the digital matched filter 2. A structure from the digitalmatched filter 2 to the correlation-point detection section 7 is thesame as that described by referring to FIG. 2.

In the tenth embodiment, the control signal corresponding to the resultof correlation-point detection output from the correlation-pointdetection section 7 is sent to the carrier control section 73. In thiscase, the carrier control section 73 performs variable control accordingto the control signal sent from the correlation-point detection section7 to increase or reduce the clock frequency of the NCO 74 until thecorrelation-point detection section 7 detects a correlation point by apeak exceeding a threshold, and maintains the frequency of the outputclock signal of the NCO 74 when the correlation-point detection section7 detects a correlation point np.

As described above, in the tenth embodiment, since the memory 71 isprovided between the A/D converter 1 and the digital matched filter 2,and data read from the memory 71 is processed at a high speed by thehigher-speed clock signal CLKa, the processing time required for thecorrelation-calculation process performed in the digital matched filter2 and the linear-addition process can be reduced. When the transfer rateis made ten times higher assuming that the digital matched filter 2 hasa margin in its hardware capability, for example, the processing time isreduced to one tenth. In searching for the carrier signal, it is notnecessary to update the received signal every time the setting of acarrier frequency is changed, but the same data stored in the memory canbe used.

In the above description, the tenth embodiment is applied to the firstembodiment. The tenth embodiment can also be applied to the second toninth embodiments.

Other Embodiments

In the above descriptions of the embodiments, the digital matchedfilter, the linear addition section, the absolute-value calculationsection, the accumulative addition section, and the correlation-pointdetection section are structured as separate pieces of hardware. All ofthese sections may be structured by one DSP. A part of each section canalso be structured by a DSP. The whole or part of each section can bestructured by software processing.

In the above descriptions of the embodiments, the present invention isapplied to a case in which a signal is received from a GPS satellite.The present invention is not limited to this case. It can also beapplied to all cases in which spreading-code synchronization capturingis applied to a signal obtained by spectrum spreading data with aspreading code.

As described above, according to the present invention, the detectionsensitivity of spreading-code synchronization for a spectrum spreadingsignal is greatly improved. Therefore, when the present invention isapplied to a GPS receiver, for example, the receiving sensitivity isimproved, and advantages, such as antenna compactness and the expansionof a receiving area, are expected.

Further, whereas a sliding correlator, which is used in a conventionalmethod, requires time, in principle, to acquire synchronization, thepresent invention greatly reduces the processing time with an effectiveuse of a high-speed DSP which employs a digital matched filter.

1. A spreading-code synchronization detection method for a spectrumspreading signal obtained by spectrum-spreading data having a bittransition period that is a multiple of one period of a spreading codewith the spreading code, comprising: a unit-periodcorrelation-calculation linear-addition step of performing a process forobtaining a linear-addition correlation-calculation result equal to avalue obtained by linear additions of results of correlationcalculations between the spectrum spreading signal and the spreadingcode, every unit period which is a multiple of one period of thespreading code and shorter than the bit transition period; anabsolute-value calculation step of calculating the absolute value of thelinear-addition correlation-calculation result obtained every unitperiod in the unit-period correlation-calculation linear-addition step;an absolute-value addition step of adding the absolute value of thelinear-addition correlation-calculation result obtained every unitperiod, obtained in the absolute-value calculation step, for a pluralityof unit periods; and a correlation-point detection step of detecting acorrelation point from a value obtained by adding the absolute values inthe absolute-value addition step; wherein the spectrum spreading signalto which the spreading-code synchronization detection is applied isstored in a memory for the plurality of unit periods for which additionsare performed in the absolute-value addition step, and the spectrumspreading signal is read from the memory at a higher speed than a speedused in writing into the memory and the unit-periodcorrelation-calculation linear-addition step and the subsequent stepsare performed to increase the speed of the process for detecting acorrelation point faster.
 2. The spreading-code synchronizationdetection method for a spectrum spreading signal according to claim 1,characterized by setting the unit period to have a time length equal tohalf the bit transition period.
 3. The spreading-code synchronizationdetection method for a spectrum spreading signal according to claim 1,characterized by using a digital matched filter to perform thecorrelation calculations between the spectrum spreading signal and thespreading code.
 4. The spreading-code synchronization detection methodfor a spectrum spreading signal according to claim 1, characterized by,in the unit-period correlation-calculation linear-addition step,linearly adding the results of correlation calculations between thespectrum spreading signal and the spreading code in the unit period. 5.The spreading-code synchronization detection method for a spectrumspreading signal according to claim 1, characterized by, in theunit-period correlation-calculation linear-addition step, in each unitperiod, synchronously adding the spectrum spreading signal at each chipphase of the spreading code in one period of the spreading code, so thatthe spectrum spreading signal is linearly added, and correlationcalculations are performed between the signal of the result of linearadditions in one period of the spreading code and the spreading code. 6.The spreading-code synchronization detection method for a spectrumspreading signal according to claim 1, characterized by, in theunit-period correlation-calculation linear-addition step, performingcorrelation calculations between data obtained by applying Fouriertransform to the spectrum spreading signal in the unit period and dataobtained by applying Fourier transform to the spreading code.
 7. Aspreading-code synchronization detection apparatus for a spectrumspreading signal obtained by spectrum-spreading data, having a bittransition period that is a multiple of one period of a spreading code,with the spreading code, comprising: unit-period correlation-calculationlinear-addition means for performing a process for obtaining alinear-addition correlation-calculation result equal to a value obtainedby linear additions of results of correlation calculations between thespectrum spreading signal and the spreading code, every unit periodwhich is a multiple of one period of the spreading code and shorter thanthe bit transition period; absolute-value calculation means forcalculating the absolute value of the linear-additioncorrelation-calculation result obtained every unit period by theunit-period correlation-calculation linear-addition means;absolute-value addition means for adding the absolute value of thelinear-addition correlation-calculation result obtained every unitperiod, obtained by the absolute-value calculation means, for aplurality of unit periods; and correlation-point detection means fordetecting a correlation point from a value obtained by adding theabsolute values in the absolute-value addition means; wherein a memoryis provided for storing the spectrum spreading signal to which thespreading-code synchronization detection is applied, for the pluralityof unit periods for which additions are performed by the absolute-valueaddition means, and the spectrum spreading signal is read from thememory at a higher speed than a speed used in storing the spectrumspread signal in the memory and the correlation calculations areperformed to increase a speed of the process for detecting a correlationpoint.
 8. The spreading-code synchronization detection apparatus for aspectrum spreading signal according to claim 7, characterized in thatthe unit period has a time length equal to half the bit transitionperiod.
 9. The spreading-code synchronization detection apparatus for aspectrum spreading signal according to claim 7, characterized in that adigital matched filter is used to perform the correlation calculationsbetween the spectrum spreading signal and the spreading code.
 10. Thespreading-code synchronization detection apparatus for a spectrumspreading signal according to claim 7, characterized in that theunit-period correlation-calculation linear-addition means comprisesmeans for linearly adding the results of correlation calculationsbetween the spectrum spreading signal and the spreading code in the unitperiod.
 11. The spreading-code synchronization detection apparatus for aspectrum spreading signal according to claim 7, characterized in thatthe unit-period correlation-calculation linear-addition means comprisesmeans for synchronously adding, in each unit period, the spectrumspreading signal at each chip phase of the spreading code in one periodof the spreading code to linearly add the spectrum spreading signal, andfor performing correlation calculations between the signal of the resultof linear additions in one period of the spreading code and thespreading code.
 12. The spreading-code synchronization detectionapparatus for a spectrum spreading signal according to claim 7,characterized in that the unit-period correlation-calculationlinear-addition means comprises means for performing correlationcalculations between data obtained by applying Fourier transform to thespectrum spreading signal in the unit period and data obtained byapplying Fourier transform to the spreading code.